Conventional delay locked loops (DLLs) include delay cells with topologies that are based on either the "current-starved" approach or on the shunt-capacitor approach, as described below and in Mark Johnson and Edwin Hudson, A Variable Delay Line PLL for CPU-Coprocessor Synchronization, IEEE Journal of Solid State Circuits, Volume 23, No. 5, pp. 1218-1223 (October 1988). FIG. 1a is a schematic circuit diagram of a conventional delay cell 100 that employs the current-starved approach. Specifically, the delay cell 100 includes an inverter 105 formed by transistors 107 and 109, a current mirror formed by the p-channel transistors 110 and 115, and n-channel transistors 120 and 125. The value of a control signal "Vcontrol" determines the current flow in the n-channel transistor 120 and the n-channel transistor 125. At lower values for Vcontrol, the current flow in the n-channel transistor 125 (or n-channel transistor 120) is low. As the value of Vcontrol increases, the current in the n-channel transistor 125 increases. A low current value provided by n-channel transistor 125 limits the value of I.sub.DS(109) which is the drain-to-source current of n-channel transistor 109. As the I.sub.DS(109) value decreases, the switching speed of inverter 105 decreases, thereby adding delay when generating the output signal V.sub.out from the input signal V.sub.in. In order to increase the delay provide by the current-starved delay element 100 to the input signal V.sub.in, the value of control signal Vcontrol is, therefore, decreased.
Conversely, as the I.sub.DS(109) value increases, the switching speed of inverter 105 increases, thereby decreasing the delay when generating the output signal V.sub.out from the input signal V.sub.in. In order to decrease the delay provided by delay element 100 to the input signal V.sub.in, the value of control signal Vcontrol is, therefore, increased.
One disadvantage of the current-starved approach is that the current values must be precise and that the devices must match in the current-starved delay element 100. Furthermore, if a short delay is to be provided to the input signal V.sub.in by the delay element 100, then the operating current typically has a high value. In addition, to compensate for process, temperature, and voltage supply variations, the sizes of the current mirrors, which consist of the p-channel transistors 110 and 115 and the n-channel transistors 120 and 125 in delay cell 100, are large in value. These characteristics disadvantageously lead to high power requirements and large die sizes. Other drawbacks for the delay cell 100 include low immunity to noise and the requirement of precise wiring to minimize noise interference. For example, conductors are preferably not crossed in the delay cell 100 in order to decrease noise interference.
FIG. 1b is a schematic circuit diagram of a conventional delay cells 150 that employs the shunt-capacitor approach. Specifically, the delay cell 150 includes an inverter 160, an n-channel transistor 165 coupled to the inverter 160 output, and a capacitor 170 coupled the n-channel transistor 165. The inverter 160 receives the input signal V.sub.in and generates the output signal V.sub.out. The delay cell 150 also includes an inverter 175, an n-channel transistor 180 coupled to the inverter 175 output, and a capacitor 185 coupled to the n-channel transistor 180.
The control signal Vcontrol controls the amount of resistance provided by the n-channel transistors 165 and 180. If, for example, the control signal Vcontrol is at a low level, then the resistance provided by the n-channel resistor 165 between the node 190 and the capacitor 170 is at a high value. An open circuit is effectively present between the node 190 and the capacitor 170, and the capacitor 170 is, therefore, not coupled as a capacitive load to the inverter 160 output. Therefore, the switching speed of inverter 160 is at a faster rate, thereby decreasing the delay of the input signal V.sub.in.
If, however, the control signal Vcontrol is at a high level, then the resistance provided by the n-channel resistor 165 between the node 190 and the capacitor 170 is at a lower value. The n-channel transistor 165, therefore, allows the node 190 to be coupled to the capacitor 170. Since the capacitor 170 acts as a capacitive load on the inverter 160 output, the switching speed of inverter 160 is at a slower rate. Thus, the delay is increased for the input signal V.sub.in.
However, the effectiveness of the delay control provided by the n-channel transistor 165 (or n-channel transistor 180) is limited for the following reason. Due to the body-effect, the threshold voltage required for turning on the n-channel transistor 165 (or n-channel transistor 180) may rise to, for example, 1.0 volts to 1.2 volts. A further disadvantage of the shunt-capacitor based delay cell in FIG. 1b is that if the capacitor 170 (or capacitor 185) is implemented by a p-type device, the capacitor 170 will be referenced to the positive voltage source VDD instead of the ground voltage VSS. As a result, when the output signal V.sub.out is switching from a low level to a high level, the capacitor 170 will be bootstrapped above the VDD level, and this condition may cause a soft breakdown for the capacitor 170. Therefore, there is a need for an improved delay cell that overcomes the aforementioned problems of conventional approaches.
Conventional delay locked loops also include phase detectors that are commonly unable to distinguish aliased signals from fundamental signals. An aliased signal occurs when the feedback clock signal in the delay locked loop is lagging a reference clock signal (received by the DLL) by more than one cycle and may often cause the DLL to lock to an inappropriate edge of the reference clock signal Refclk. Therefore, there is a need for a delay locked loop that can compensate for aliased signals, thereby leading to improved DLL performance.